Known are voltage converters (or, in a similar way, regulators or power supplies) having a galvanic insulation between an input voltage and a regulated output voltage, having a desired value, wherein the galvanic insulation is obtained via a transformer having a primary winding receiving the input voltage, and a secondary winding supplying the regulated output voltage. Generally used are two techniques for controlling these voltage converters, which envisage a feedback either on the secondary side or on the primary side of the transformer. In the first case, a feedback voltage is taken directly on a secondary winding of the transformer, in parallel to the output, and sent to a regulation circuit via an optocoupler device, so as to maintain the galvanic insulation. In the second case, the feedback voltage is generally taken on an auxiliary winding, purposely provided on the primary side of the transformer. The feedback on the primary side makes it possible to avoid the use of external insulation devices (for example, additional optocouplers or transformers), but entails higher levels of consumption and hence a degradation of the efficiency of regulation.
A wide range of control techniques has been proposed for implementing an efficient voltage regulation with feedback from the primary winding, but so far none of these has proven altogether satisfactory.
In particular, the use has been proposed of a purposely provided sample-and-hold device for sampling the feedback voltage on the auxiliary winding at the end of demagnetization of the transformer, i.e., when the value of this voltage corresponds to the value of the output voltage, being, in a know way, a faithful replica thereof.
In detail, and as is shown in FIG. 1, a voltage converter 1, of an isolated flyback type with control of the peak current and feedback on the primary winding, has a first input terminal IN1 and a second input terminal IN2, which are designed to receive an input voltage Vin, for example from a voltage generator 2, and a first output terminal and a second output terminal OUT2, between which an output capacitor 3 is OUT1 coupled and an output voltage Vout with regulated value is present. The voltage converter 1 supplies to a load an output current Iout.
The voltage converter/comprises a transformer 4, having a primary side and a secondary side electrically isolated from the primary side, and having a primary winding 5, a secondary winding 6, and an auxiliary winding 7 (the latter positioned on the primary side of the transformer 4). For example, the transformer 4 has a turn ratio N between the primary winding 5 and the secondary winding 6, and a unit turn ratio between the secondary winding 6 and the auxiliary winding 7. The primary winding 5 has a first terminal, which is coupled to the first input terminal IN1, and a second terminal, which is coupled to a control switch 8, which can be actuated for controlling PWM operation of the voltage converter 1. The secondary winding 6 has a respective first terminal, which is coupled to the first output terminal OUT1, via the interposition of a first rectifier diode 9, and a respective second terminal, which is coupled to the second output terminal OUT2. The auxiliary winding 7 has a respective first terminal, on which an auxiliary voltage Vaus is present and which is coupled to a resistive divider 10, and a respective second terminal, which is coupled to a reference potential. FIG. 1 shows the magnetization inductance Lm of the primary winding of the transformer 4, coupled across the primary winding 5, and the leakage inductance Lpe of the same primary winding 5, coupled to the first input terminal IN1.
The control switch 8, for example a power MOS transistor, has a first conduction terminal, which is coupled to the primary winding 5, a second conduction terminal, which is coupled to the reference potential, via the interposition of a sense resistor 11, and a control terminal, which is coupled to a control circuit 12, designed to control PWM operation of the voltage converter 1.
The resistive divider 10 includes a first resistor 13 and a second resistor 14, which are coupled in series between the first terminal of the auxiliary winding 7 and the reference potential and define an intermediate node 15 having a feedback voltage Vfb (proportional to the auxiliary voltage Vaus).
The voltage converter 1 further comprises a self-supply capacitor 16, which is coupled to the auxiliary winding 6 via the interposition of a second rectifier diode 17 and is designed to supply, in a known way, a self-supply voltage Vcc to the control circuit 12 during the demagnetization of the transformer 4.
In detail, the control circuit 12 has a first input, which is coupled to the intermediate node 15 and receives the feedback voltage Vfb, a second input, which is coupled to the sense resistor 11 and receives a sense voltage Vs (proportional to the current circulating in the primary winding 5), and an output, which is coupled to the control terminal of the control switch 8 and supplies a driving signal DR.
The control circuit 12 comprises: a sampling stage 20, which is coupled to the intermediate node 15 and supplies at output a sampled signal FB, which is the result of sample and hold (for example, performed at each switching cycle) of the feedback voltage Vfb at the end of the demagnetization step; an error-amplifier stage 22, having a first input terminal, which is coupled to the output of the sampling stage 20 and receives the sampled signal FB, a second input terminal, which is coupled to a first reference generator 23 and receives a first reference signal Vref, the value of which is a function of a desired value of the regulated output voltage Vout, and an output terminal, which is coupled to an external compensation network 24 (represented schematically in FIG. 1 by a load impedance). On the output terminal of the error-amplifier stage 22 a control signal Vcon (a voltage signal) is present.
The control circuit 12 further comprises: a first comparator 27, designed to compare the control signal Vcon with the sense voltage Vs; a driving block 28, which is cascaded with the first comparator 27 and is designed to generate the driving signal DR as a function of the result of the aforesaid comparison (comparison signal drv_off) and of a driving signal drv_on received at input from a clock generator 29.
The voltage converter 1 further comprises a snubber network 30, of a passive type, coupled across the primary winding 5 of the transformer 4. The snubber network 30 comprises a recirculation diode 31, and a clamp resistor 32 coupled in parallel to a clamp capacitor 33 between the first input terminal IN1 of the voltage converter/and the second terminal of the primary winding 5 via the interposition of the recirculation diode 31.
General operation of the voltage converter/illustrated previously is now briefly described.
Due to the absence of an optocoupler between the secondary side of the transformer 4 and the control circuit 12, the value of the output voltage Vout is read from the auxiliary winding 7, via the resistive divider 10 upstream of the second rectifier diode 17. In the ideal case of absence of leakage inductances and of parasitic resistances of the transformer 4 and of the wires, and assuming the voltage drop on the first rectifier diode 9 negligible, the auxiliary voltage Vaus taken on the auxiliary winding 7 is proportional to the output voltage Vout during the period in which, between one switching cycle and the next, the first rectifier diode 9 is in conduction, basically for the entire duration of demagnetization of the transformer 4. In actual fact, on account of the leakage inductances of the transformer and of the equivalent resistance on the secondary winding of the transformer 4, superimposed on the useful signal of the auxiliary voltage Vaus is a damped oscillation, which causes the auxiliary voltage Vaus to be a faithful replica, but for the turn ratio of the transformer 4, of the output voltage Vout only at the instant in which the demagnetization of the transformer 4 is concluded. In fact, in this instant of time the current on the secondary winding is zero, and hence the equivalent resistance on the secondary winding has no effect, and moreover the oscillations due to the leakage inductances have ended (assuming that the demagnetization time is sufficiently long).
The plot of the output signal Vout and of the auxiliary voltage Vaus is shown in FIG. 2a, in which the demagnetization period is designated by Tdem. FIG. 2b shows the corresponding plot of the demagnetization current Idem, which becomes zero at the end of the demagnetization period Tdem.
The sampling stage 20 is consequently configured to sample the feedback voltage Vfb at the instant of demagnetization of the transformer 4, so that the sampled signal FB coincides, but for the turn ratio of the transformer 4 and the dividing ratio of the resistive divider 10, with the output voltage Vout.
The difference between the first reference signal Vref, which represents the value of the output voltage to be regulated, and the sampled signal FB constitutes the error signal at input to the error-amplifier stage 22. In addition, the control signal Vcon at output from the error-amplifier stage 22 determines the peak of current on the primary winding, and hence the switching time of the control switch 8 (in PWM mode). In particular, the driving block 28 supplies to the control terminal of the control switch 8 the driving signal DR, and charges the magnetization inductance Lm of the transformer 4 with an energy proportional to the square of the aforesaid peak current.
An operating condition in which the driving signal DR has minimum duty cycle and frequency values is known as “burst-mode condition” (or low-consumption condition). This operating condition arises in the presence of a very low output load. In order to reduce the power consumption of the voltage converter 1, the driving block 28 drives the control switch 8 with a switching frequency much lower than the one used in conditions of normal load and ordinary operation (for example, with a frequency of 1 kHz, instead of 50 kHz). The switching pulses supplied to the control terminal of the control switch 8 are hence spaced further apart in time.
The function of the snubber network 30 is that of limiting the voltage overshoots on the conduction terminal of the control switch 8 coupled to the primary winding of the transformer 4, after its turn-off. The energy that is stored in the leakage inductance Lpe during the turn-on phase of the control switch 8 is in fact transferred to and dissipated in the snubber network 30 during the turn-off phase.
One of the limits of the system for regulation of the output voltage Vout described above is represented by the fact that, especially in the burst-mode condition, the presence of the snubber network 30 on the primary winding 5 may affect the reading of the output voltage Vout and consequently jeopardize regulation of the same output voltage, unless the choice is made to sacrifice the efficiency of the voltage converter 1.